非常经典的关于LLC的杨波博士论文chapter5

Bo Yang Chapter 5. Improvements of LLC resonant converter 142 Chapter 5 Improvements of LLC Resonant Converter From previous chapter, the characteristic and design of LLC resonant converter were discussed. In this chapter, two improvements for LLC resonant converter will be investigated: integrated magnetic design and over load protection. 5.1 Magnetic design for LLC Resonant Converter From previous discussion, the power stage could be designed according to the given specifications. The outcome of the design is the desired values for the components. For these components, power devices and capacitors are obtained from manufactures, which already reflect the state of the art technology. Within all these components, magnetic is the one need to be physically designed and built by power electronics researcher. In this part, the design of magnetic component for LLC resonant converter will be discussed. 5.1.1 Discrete design and issues For a LLC resonant converter, the magnetic components need to be designed are shown in Figure 5.1. There are three magnetic components: Lr, Lm and transformer T. From the configuration of Lm and transformer T, it is easy to build Bo Yang Chapter 5. Improvements of LLC resonant converter 143 Lm as the magnetizing inductance of transformer. So in fact, we are trying to build one resonant inductor and one transformer with magnetizing inductance. Figure 5.1 Magnetic structure for LLC resonant converter There are several ways to build them. One is using discrete components, with one magnetic core to build the resonant inductor and one magnetic core to build the transformer and magnetizing inductor Lm. The benefit of this method is that the design procedure is well established. Next, a discrete design is presented and simulation result is showed to provide a reference for later integrated magnetic designs. For LLC resonant converter, the resonant inductor Lr has pure AC current through it, so we use soft ferrite core for both inductor and transformer. Figure 5.2 shows the discrete design of the magnetic for LLC resonant converter. Two U cores were used to build the resonant inductor and gapped transformer. Fig.6 shows the simulation results of flux density in the core. For Bo Yang Chapter 5. Improvements of LLC resonant converter 144 each U core, the cross-section area is 116.5mm2. Design result: nl=12, np: ns: ns=16:4:4, gap1=1.45mm and gap2=0.6mm. (a) (b) Figure 5.2 Discrete magnetic design (a) schematic (b) physical structure (a) Inductor (b) Transformer Figure 5.3 Flux density simulation result (a) Inductor, and (b) Transformer Bo Yang Chapter 5. Improvements of LLC resonant converter 145 Figure 5.3 shows the flux density in each core at 400V input with switching frequency at 200kHz. As seen in the graph, the flux densities in both cores are pretty high. Both cores with high flux density excitation will contribute to the total core loss. For high frequency, core loss is a major limitation on pushing to higher frequency and smaller size. Figure 5.4 shows the peak-to-peak flux density for each core with different input voltage. At low input voltage, the flux density will increase, but it is not critical because of short operating time. (a) Inductor (b) Transformer Figure 5.4 Peak to peak flux density under different input voltage at full load Bo Yang Chapter 5. Improvements of LLC resonant converter 146 The drawbacks of this method are: 1. Two magnetic cores are needed, which results in more components count and connections, 2. High magnetic loss caused by high flux ripple in magnetic structure, 3. Large footprint is needed for the whole structure. In recent years, integrated magnetic has been investigated for many different applications. For asymmetrical half bridge with current doubler, all the magnetic components could be integrated into one magnetic structure with integrated magnetic concept [C1][C5]. In this part, the integrated magnetic structure will be discussed for LLC resonant converter. It integrated all magnetic components into one magnetic core. Through magnetic integration, the component count and footprint are reduced, the connections is also reduced. With proper design; flux ripple cancellation can be achieved, which can reduced the magnetic loss, and reduce the magnetic core size. In the next part, the integrated magnetic designs for LLC resonant converter will be discussed and compared. 5.1.2 Integrated magnetic design 5.1.2.1 Real transformer with leakage and magnetizing inductance First structure is just use one transformer and uses the leakage inductance as resonant inductor. The configuration of magnetic components for LLC resonant converter is exactly the same as a real transformer with magnetizing inductance Bo Yang Chapter 5. Improvements of LLC resonant converter 147 and leakage inductance. It is natural to think about using one real transformer to get all the needed components. The issues with structure are: 1. The leakage inductance cannot be accurately controlled which will determine the operating point of the converter, 2. When we build Lr this way, the leakage inductance will not only exist on primary side, it will also exist on secondary side of the transformer. So the result get from real transformer will be as in Figure 5.5. Llp and Lls have similar value when transferred to same side of the transformer. Figure 5.5 Structure with real transformer Figure 5.6 Desired magnetic components configuration Bo Yang Chapter 5. Improvements of LLC resonant converter 148 Figure 5.7 Magnetic components configuration from real transformer (a) (b) Figure 5.8 Voltage stress of output diodes D1 D2 (a): desired structure (b) real transformer When the leakage inductance exists on secondary side, it will increase the voltage stress on secondary rectifier diode. This requires us to use higher voltage rating diode, which will increase the conduction loss of the output rectifier. Figure 5.8 shows the simulate waveforms of secondary diodes voltage stress with Bo Yang Chapter 5. Improvements of LLC resonant converter 149 magnetic structure in Figure 5.6 and Figure 5.7. We can see that with inductor on the secondary side, the voltage stress of the diodes is much higher. From above discussion, we can see that the desired magnetic structure will need to provide accurate control of Lr and Lm, at that same time, minimize the inductance on secondary side, which could not be achieved with just a transformer with leakage and magnetizing inductance. Next more complex integrated magnetic structure will be investigated. 5.1.2.2 Integrated magnetic design A From discrete design, just combine them together with an EE core, we will be able to integrate the two components into one magnetic component as shown in Figure 5.9. Figure 5.9 Integrated Magnetic Designs A E42/21/20 core is used. The cross-section area of is 233mm2. For the outer legs, they have same cross-section area as discrete design. Turn number nl, np and ns is the same as in discrete design. For this design, the inductor and transformer Bo Yang Chapter 5. Improvements of LLC resonant converter 150 design is decoupled. Discrete design procedure still can be used. Figure 5.10 shows the simulation result of for this structure. Figure 5.10 Flux density simulation result for Design A It can be seen from the simulation result: for inductor and transformer leg, the flux density is the same as discrete design. But for center leg, the flux density is much smaller than discrete case. This will greatly reduce the magnetic loss in the big part of the magnetic component. Figure 5.11 Center leg flux density for different input voltage Bo Yang Chapter 5. Improvements of LLC resonant converter 151 Figure 5.11 shows the center leg flux density for whole input voltage range. Compare with discrete design, the flux density is only half of the transformer leg and much smaller than inductor leg within all input voltage range. The problem for this structure is the gapping. In this structure, we are using E cores. The air gap is on two outer legs while there is no air gap on center leg. This structure is not good in several aspects: first, this core structure is not a standard. The standard core normally has air gap on the center leg or no air gap at all. Second, it is not a mechanical stable structure, very accurate gap filling need to be provided. Otherwise, the accuracy of the components value will be impacted. Also, when force is applied which happens when the converter is working, the core tends to vibrate. This vibration will cause broken of the core. A desired core structure will have air gap on center leg or same air gap for all three legs. Following part will try to establish an electrical circuit model for a general integrate magnetic structure. From the model, we can investigate new core structures. 5.1.2.3 Extraction of Common Structure for Integrated Magnetic In the past, lot of research was done on integrated magnetic design for power converters. Review those paper, we can find that most of them are based on EE core structure or three legs structure. The difference between different designs is the placement of windings and air gaps. Bo Yang Chapter 5. Improvements of LLC resonant converter 152 In this part of the paper, the general circuit model of an EE core with four windings is used as a general structure as shown in Figure 5.12. There are air gaps on each leg. This is a very commonly used structure, many integrated magnetic design for PWM converter also used this structure with some change on the air gap or winding placement [C5]. The reason of choosing this structure for LLC resonant converter is as following: To integrate two magnetic components, usually we need three magnetic paths. In the LLC resonant converter, although we have three magnetic components, Lm and transformer T can be build with an air-gapped transformer. So in fact we need integrated two magnetic components: series resonant inductor Lr and gapped transformer T. An EE core structure will be a reasonable choice. Figure 5.12 general magnetic structures for Integrated magnetic The model is derived through duality modeling method [E4]. Through this method, we can get the electrical circuit model of a physical magnetic structure. All the components in the model are related to the physical structure of the magnetic structure. Figure 5.13 shows the reluctance model of magnetic structure Bo Yang Chapter 5. Improvements of LLC resonant converter 153 shown in Figure 5.12. Figure 5.14shows electrical circuit model form this structure. In the structure, we have two sets of ideal transformer and three inductors. Figure 5.13 Reluctance model of general integrated magnetic structure For the two ideal transformers, they have same turns ratio as in real physical structure. For the three inductors, they are correspond to each air gap and reflected to first winding n1. They can also be reflected to other windings as necessary. The value of each inductors are as following: Figure 5.14 Circuit model of general integrated magnetic structure Base on this circuit model, we will investigate more integrated magnetic structures. Bo Yang Chapter 5. Improvements of LLC resonant converter 154 5.1.2.4 Integrated magnetic design B for LLC resonant converter As discussed in structure A, the air gapping for structure A is not easy to implement. In this part, we will investigate structure with same air gap for all three legs and same winding structure as shown in Figure 5.15. Figure 5.15 Integrated Magnetic Designs B The electrical model of this structure can be easily got from general structure. Compare this structure with general structure; design B has only one winding on left side leg. By simplify the general model we can get following circuit model of design B as shown in Figure 5.16. Figure 5.16 Electrical circuit model of integrated magnetic structure B Base on the electrical circuit model of the structure, next terminal 2 and 3 are connected, which gives following circuit model. Bo Yang Chapter 5. Improvements of LLC resonant converter 155 Figure 5.17 Electrical model of connecting dot-marked terminal with unmarked terminal From circuit model in Figure 5.17, write the input current and voltage equations and solve them, then we can get the equivalent circuit of the structure. For this circuit, it has two modes. One mode is n3 is connected to output voltage. During this mode, the energy is transferred from primary to output. During the other mode, both secondary windings n3 are not connecting. We will derive the equivalent circuit for these two modes separately. (mode a) (mode b) Figure 5.18 Two operation modes for LLC resonant converter For operation mode (a), we can get following equations: Bo Yang Chapter 5. Improvements of LLC resonant converter 156 in 1 v=v n1 n2 dt diL1 1+ in 0 v=v n1 n2v dt diL0 11 ++ Vo n3 n1=v1 ini=ii 10 + From above equations, we can get the relationship of input voltage, input current and output voltage as following: ) L0L1 L1n1(n2 n3 Vo dt di L0L1 L0L1=vin in + ++ + ⋅ 1 From this equation, we can get the equivalent circuit during this mode as in Figure 5.19. Figure 5.19 Equivalent-circuit for mode (a) In this circuit, Lr, Lm and na are as following: L0L1 L0L1=Lr + ⋅ Bo Yang Chapter 5. Improvements of LLC resonant converter 157 L0L1 L1n1n2=na + + To find out Lm, we need to analyze mode (b). Same as analysis for mode (a), we can get following equations for Lm: L0L2L1 L0L1 n1 naL2=Lm 2 ++ + ⋅⋅ 2 From the equivalent circuit, derive the relationship between terminals; the equivalent circuit above can be simplified into the equivalent circuit, which is the structure we desired. The relationship between resonant inductor, magnetizing inductance and transformer turns ratio is shown also. Base on these equations, the structure can be designed. Following is an example of design: turns ratio 12:3, Lm=14u and Lm = 60uH. The relationship of above equations could be drawn in Figure 5.20. For given turns ratio, there are many different ways to choose n1 and n2 to get the desired na, for example, n1=n2=9, n1=6 and n2=10. The other constrain will be the desired Lm. For this case, the Lm is 4.5 times Lr. To get this Lm, the n2 need to be choosing as 10. Bo Yang Chapter 5. Improvements of LLC resonant converter 158 (a) (b) Figure 5.20 Design curves for integrated magnetic structure B for LLC converter From above discussion, n1=6, n2=10 and n3=3 give us turns ratio 12:3, Lm/Lr = 4.5. Next step will be design the air gap, we knows n1 and L1 value. Follow tradition inductor design equations, the air gap can be designed. Here Lr = 14uH, from the structure it can be seen that: L1 = L3 = 0.5 L2. From the relationship above, it can be calculated that we need L1=21uH to give us equivalent Lr=14uH. With the core cross-section area and turns given, the gap can be easily derived. In this part, the detailed information of the magnetic is described. For this converter, the core used is EE56/24/19 from Phillips. The core material is 3F3. Two outer legs are used to wind the windings. Air gap is 0.55mm for all legs. Primary windings are built with 8 strands of AWG#27 wires. Secondary side uses 5mil X 0.9inch copper foil. Bo Yang Chapter 5. Improvements of LLC resonant converter 159 Figure 5.21 shows the simulated flux density on each of the legs. From simulation result we can see that the flux density on center leg is greatly reduced. So with this integrated magnetic structure, we can reduce the core loss greatly. Also, with this structure, the air gap is the same for all legs, which is easier to manufacture and doesn’t have mechanical problem. Figure 5.21 Flux density in each leg for integrated magnetic structure B 5.1.3 Test Result In this part, the test result of integrated magnetic structure B is tested. It is compared with a discrete design. The test efficiency of integrated magnetic and discrete magnetic is shown in Figure 5.22. Because of flux ripple cancellation effect and less turns number, although the size of the magnetic components is reduced, the efficiency is almost the same for these two designs. In Figure 5.23, the sizes of these two designs were compared. With integrated magnetic, the footprint of the magnetic components could be reduced by almost 30%. Bo Yang Chapter 5. Improvements of LLC resonant converter 160 Figure 5.22 Efficiency comparison of integrated and discrete magnetic design for LLC converter Figure 5.23 Magnetic size comparison of discrete and integrated magnetic 5.1.4 Summary In this part, the magnetic design for LLC resonant converter is discussed. Discrete design and three method of integrated design were investigated. For discrete design, the footprint is pretty large. Also, there is no flux ripple cancellation effect; the magnetic loss is high in discrete design too. With real transformer, the magnetic components could be built with one magnetic structure. The problem is difficult to control the leakage inductance. Another integrated Bo Yang Chapter 5. Improvements of LLC resonant converter 161 magnetic structure is to integrate the two U cores used to build discrete magnetic. With this method, the problem is the mechanical structure is not a stable structure. To improve this structure, a general integrated magnetic structure is developed. With the model, another integrated magnetic structure is developed with same air gap on all legs. With this magnetic structure, the manufacture is easy. There is no mechanical problem. Also, flux ripple cancellation could be achieved with this structure. Compare with di